Converter topologies with multiple windings

ABSTRACT

Converter topologies in which two separate switching transistors and two capacitors are used on the input side. Preferably the two transistors are switched alternately, to alternately pull down different nodes in an inductor-capacitor chain. Two capacitors are interposed in series between an input inductor on the input and a transformer primary winding. Preferably the two transistors are connected with their parasitic diodes in opposite senses, so that one can source current from a first node to ground when off, and the other can sink current from a second node to ground when off.

BACKGROUND AND SUMMARY OF THE INVENTION

The present invention relates to "off-line" power converters in portable computers.

Background: Switched-Mode Power Conversion

One of the recurring problems in electronic system design is the need to derive power from an externally determined source voltage to operate at a different voltage. In particular, "off-line" converters address the frequent need to efficiently translate the relatively high voltages of power mains (120V or 240V) into the lower voltages which are actually used by systems which include integrated circuits (e.g. 5V, 3.3V, or less). In portable systems, small volume (for a given power capability) and high efficiency are important goals for converters and power supplies. Power converter design has long been a very active area of engineering effort, and numerous circuit topologies have been proposed. See, e.g., Pressman, SWITCHING POWER SUPPLY DESIGN (1991); the 3 volumes of Middlebrook and Cuk, ADVANCES IN SWITCHED-MODE POWER CONVERSION (2.ed. 1983); and all of the biennial UNITRODE POWER SUPPLY DESIGN SEMINAR HANDBOOKS.

Background: Flyback Converters

Flyback converters are a class of power converter topology which is common in off-line adapters. In this configuration a switching transistor switches the primary of a transformer. Each of the secondary windings of the transformer is connected through a diode (the "flyback" diode) to an output capacitor. While the switching transistor is on the current through the primary increases (storing energy in the transformer), and when the switch is opened, the stored energy is discharged at the transformer secondary through the flyback diode. Feedback is provided through a pulse-width modulation circuit which varies the switching waveform as the load impedance varies. The duty cycle and the turns ratio of the transformer are chosen to provide the desired output voltage range.

Innovative Converter Topologies with Decoupled Transformers

The present application describes new converter topologies in which two independent and decoupled power transformers are used in a single integrated magnetic structure.

Innovative Converter Topologies with Anti-Parallel Switching Transistors on Primary Side

The present application also describes new converter topologies in which two separate switching transistors and two capacitors are used on the input side. Preferably the two transistors are switched alternately, to alternately pull down different nodes in an inductor-capacitor chain. Two capacitors are interposed in series between an input inductor on the input and a transformer primary winding. Preferably the two transistors are connected with their parasitic diodes in opposite senses, so that one can source current from a first node to ground when off, and the other can sink current from a second node to ground when off.

This permits ripple to be suppressed at the input, and also provides a clamping effect to suppress transient overvoltages. Thus this class of circuits solves a longstanding problem in isolated SEPIC converters.

BRIEF DESCRIPTION OF THE DRAWING

The disclosed inventions will be described with reference to the accompanying drawings, which show important sample embodiments of the invention and which are incorporated in the specification hereof by reference, wherein:

FIG. 1 shows an innovative integrated magnetic structure according to the presently preferred embodiment.

FIG. 2A shows a first sample power converter circuit topology, of the SEPIC type, which is advantageously implemented using the magnetic structure of FIG. 1.

FIG. 2B shows a second sample power converter circuit topology which is advantageously implemented using the magnetic structure of FIG. 1.

FIG. 3A shows the innovative "segregated flyback" power converter circuit topology, which can advantageously implemented using the magnetic structure of FIG. 1, and FIG. 3B is a timing diagram which shows the operation of the circuit of FIG. 3A.

FIG. 4A shows the innovative "dual regenerative flyback" power converter circuit topology, which can advantageously implemented using the magnetic structure of FIG. 1, and FIG. 4B is a timing diagram which the operation of the circuit of FIG. 4A.

FIG. 4C shows another dual regenerative flyback converter circuit.

FIG. 5A is a plan view of the magnetic structure of FIG. 1, according to the presently preferred embodiment. FIG. 5B is a plan view of an alternative embodiment of the magnetic structure of FIG. 1. FIG. 5C is a plan view of another alternative embodiment of the magnetic structure of FIG. 1. FIG. 5D is a plan view of yet another alternative embodiment of the magnetic structure of FIG. 1.

FIG. 6A shows an alternative integrated magnetic structure with asymmetrical gapping, and FIG. 6B is a magnetic circuit diagram of the structure of FIG. 6A.

FIG. 7 shows a prior art planar magnetic structure.

FIG. 8 shows a block diagram of a portable computer system according to the presently preferred embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The numerous innovative teachings of the present application will be described with particular reference to the presently preferred embodiment. However, it should be understood that this class of embodiments provides only a few examples of the many advantageous uses of the innovative teachings herein. In general, statements made in the specification of the present application do not necessarily delimit any of the various claimed inventions. Moreover, some statements may apply to some inventive features but not to others.

FIG. 1 shows an innovative integrated magnetic structure according to the presently preferred embodiment. This includes a C-shaped core piece 110, and a flat lid 120 which is assembled to the core piece 110. The core piece 110 and lid 120 are both made of high-permeability material, which in a sample embodiment is "P" material from Magnetic Inc. (This material has an initial permeability of approximately 2500.) In the example shown the cross section (in the plane of the page) of both plates, and both poles, is 0.10 inches. The width of the structure (i.e. the front-to-back dimension, which is normal to the page in the orientation illustrated) is 0.50 inches in the presently preferred embodiment. The height of each of the two legs of the `C` shaped core is 0.2 inches overall. The total assembled height is approximately 0.3 inches.

The air gap of the two legs is defined by a teflon wrap on the planar lid 120, and is about 0.005 inches in the presently preferred embodiment. With this structure, it was experimentally determined that the inductive coupling coefficient k between two 20-turn planartype coils placed on the two poles was less than 0.1 at 50 kHz.

FIG. 1 shows a sample embodiment of the innovative planar magnetic structure. A single molded C-core body 110 of high-permeability material has only two poles p₁ and P₂. (The view shown is cross-sectional, but the body 110 has the same cross-section all the way through.) One winding (or winding stack) W_(t) is wound around pole p₁, and the flux generated by this winding W₁, recirculates through the lid 120 (which is also molded of the same high-permeability material) and the open space 130. Another winding (or winding stack) W₂ is wound around pole p₂, and the flux generated by this winding W₂ recirculates through the lid 120 (which is also molded of the same high-permeability material) and the open space 130. Note that the two windings W₁ and W₂ drive separate flux paths φ₁ and φ₂ respectively. The two flux paths both include a significant part of the width w of the open space 130 which laterally separates the two windings W₁ and W₂. In this embodiment air gaps G₁ and G₂ (which are not actually filled with air, but by a polymer film) are included in the two poles p₁ and p₂. Note that the reluctance of the flux paths φ₁ and φ₂ decreases as W is increased, and decreases rapidly as the height l of the open space 130 is decreased. Gap G₁ helps prevent flux path φ₂ from being shunted by flux conduction through pole p₁, and gap G₂ helps prevent flux path φ₁ from being shunted by flux conduction through pole p₂.

This structure permits more power to be transferred through a core of a given size. For a flyback application, the continuous power which can be transferred is:

    P=(V.sub.in Iδ)/2

where

    δ=((NV.sub.out)/(V.sub.in +NV.sub.out)).

So for a given Vin and Vout, power is a function of peak current only, i.e. δ does not vary with inductance. So the maximum power is

    Pmax=((V.sub.in *δ)/2)*I.sub.max =((V.sub.in *δ)/2)((B.sub.sat l)/Nμ).

But because the two halves of the C-core are independent (the two halves have effectively independent flux return paths), each does not contribute to the saturation of the other, so P_(c) =2*P_(e).

FIG. 5A is a plan view of the magnetic structure of FIG. 1, according to the presently preferred embodiment. In this view the core piece is shown from the side on which the poles are located, and the lid is not shown. Preferably (but not necessarily) the lid 120 has the same outer dimensions as the C-core 110.

FIG. 5B is a plan view of an alternative embodiment of the magnetic structure of FIG. 1. For comparison, the dimensions of the pole pieces and coils may be assumed to be exactly the same in the four embodiments of FIGS. 5A-5D. In this embodiment the modified core 110' extends beyond the poles p₁ and p₂ in one direction. Preferably (but not necessarily) the lid 120 has the same outer dimensions as the C-core 110'.

FIG. 5C is a plan view of another alternative embodiment of the magnetic structure of FIG. 1. In this embodiment the modified core 110" extends beyond the poles p₁ and p₂ in one direction (different from that used in FIG. 5B). Preferably (but not necessarily) the lid 120 has the same outer dimensions as the C-core 110 ".

FIG. 5D is a plan view of yet another alternative embodiment of the magnetic structure of FIG. 1. In this embodiment the modified core 110'" extends beyond the poles p₁ and p₂ in two directions. Preferably (but not necessarily) the lid 120 has the same outer dimensions as the C-core 110'".

Asymmetric-Coupling Structures

A further class of alternative embodiments uses asymmetric gapping for the two flux paths. If a "planar" magnetic structure has an air gap G₁ atop pole p₁ but no gap G₂ atop pole p₂, then flux generated by a winding W₂ on pole p₂ will largely be returned through the air return path described above. (More precisely, one gap includes a Teflon spacer, whereas the other gap is typically replaced by a butt joint. A butt joint will add some reluctance, but less than that of an intentionally-added air gap.) Thus an AC signal applied to the winding W₂ on pole p₂ will not be coupled strongly into the winding W₁ on pole p₁, i.e. the coupling from W₂ to W₁ is very weak. By contrast, for flux generated by the winding W₁ on pole p₁, the air return path is shunted by a lower-reluctance circuit through the magnetic material (passing through pole p₂), so this flux will be coupled to winding W2. Thus an AC signal applied to the winding W₁ on pole p₁ will not be coupled significantly into the winding W₂ on pole p₂, i.e. the coupling from W₁ to W₂ is stronger than the coupling from W₂ to W₁. This asymmetry in coupling can be advantageously exploited in various ways, e.g. for ripple-steering.

FIG. 6A shows an alternative integrated magnetic structure with asymmetrical gapping, and FIG. 6B is a magnetic circuit diagram of the structure of FIG. 6A. This magnetic circuit diagram shows two mmf sources F_(w1) and F_(w2) (provided by the two windings W₁ and W₂), and three reluctances R_(G1), R_(G2), and R_(rt) (provided respectively by gap G1, gap G2, and the air return path 130). In the embodiment of FIG. 1, the magnitude of either gap reluctance R_(G1) or R_(G2) is typically larger than the magnitude of the reluctance R_(rt) of the air return path; however, in the embodiment of FIG. 6A, the gap reluctance of one gap R_(G2) is much smaller than the magnitude of the reluctance R_(rt) of the air return path. This results in the asymmetric coupling described above.

Alternatively, a lesser degree of asymmetry can be achieved by making one of the gaps G₁ and G₂ nonzero, but significantly smaller than the other gap.

Power Converter Topologies

The double-circuit integrated magnetic structures make certain circuit topologies more attractive. Several such examples will now be listed, but of course this does not preclude others.

Modified SEPIC-Type Converter

As a first example of use of the magnetic structure of FIG. 1, FIG. 2A shows a SEPIC style converter circuit, in which an input inductor L₀ on the input is completely decoupled from the main transformer L_(p) /L_(s). A switch SW modulates the current through the input inductor, and a capacitor C_(i) is interposed between the input inductor and the transformer primary L_(p). In this circuit the input inductor L₀ is preferably placed on one leg p₁ of a magnetic structure like that of FIG. 1, and the transformer primary and secondary coils L_(p) /L_(s) are both stacked on the second leg p₂ of the magnetic structure.

In the structure of FIG. 2A, a conventional output diode D_(F) and output capacitor C_(out) provide an output voltage V_(o), which is controlled by the turns ratio between the primary and secondary, and also by the duty cycle of the switch SW. As with other SEPIC-type converters, the structure in FIG. 2A steers ripple from the input side into the transformer primary L_(p), i.e. no ripple appears on the input voltage V_(in). A normal SEPIC-type converter can use a single magnetic structure, but the use of segregated magnetics provides reduces size for a given power output.

Note that no snubbing circuit is shown in FIG. 2A, and indeed no snubbing circuit is preferably used.

Compact Decoupled-Dual-Transformer Converter

Another circuit topology which can be used with the present invention (but which is not as advantageous as the embodiment of FIG. 3A in some respects) is the decoupled-dual-transformer circuit topology shown in FIG. 2B. One transformer is formed by coils L₂ and L₃, which are tightly coupled together, and the other transformer is formed by coils L₁ and L₄, which are also tightly coupled together. A capacitor C₂ is interposed between windings L₁ and L₂, and a capacitor C₃ is interposed between windings L₃ and L₄. Switch SW1' modulates the voltage across inductor L₁. Capacitors C₁ and C₄ provide input and output filtering respectively.

This circuit has some resemblance to an isolated (Cuk converter circuit, but has different magnetic connections and operates differently. In an isolated Cuk converter, both L1 and L4 are magnetically coupled to the L2/L3 pair. (Typically the L₂ and L₃ coils would be wound on the center leg of an E-core magnetic structure, L₁ would be wound on one outer leg of the same E-core magnetic structure, and L₄ on the other leg.) See ADVANCES IN SWITCHED-MODE POWER CONVERSION (2.ed. Middlebrook and Cuk 1983), which is hereby incorporated by reference. Thus a correct circuit diagram of an isolated (Cuk converter would show a magnetic coupling path between the L₁ /L₄ transformer and the L₂ /L₃ transformer, but this magnetic coupling path is absent in the circuit of FIG. 2B.

By contrast, the circuit topology of FIG. 2B is implemented with a segregated magnetics structure like that of FIG. 1, and hence the L₁ /L₄ transformer is decoupled from the L₂ /L₃ transformer. In one sample implementation the L₁ and L₄ inductors are tightly coupled together on one pole p₁, and the L₂ and L₃ windings are tightly coupled together on the other pole p₂. However, without some degree of coupling between these two pairs, ripple cancellation will not be achieved at both input and output sides. (Ripple reduction can be achieved on one side, but not both.) Thus the topology of FIG. 2B sacrifices some of the advantages of the isolated (Cuk converter, for the sake of reduced size and weight.

In a further alternative embodiment, the height l of the planar structure of FIG. 1 is therefore increased, to increase the reluctance of the return paths φ₁ and φ₂ and introduce some additional coupling between the two transformers of this single magnetic circuit. (Thus this alternative is a hybrid between the circuit of FIG. 2B and an isolated Cuk converter.) However, this degrades the low form factor and low volume which is preferred for magnetic circuits.

Segregated Flyback Converter

Another circuit topology which can use the magnetic structure of FIG. 1 is shown in FIG. 3A. (This topology is also believed to be independently innovative, apart from its use with the magnetic structure of FIG. 1.) This circuit topology uses two transformers with the same turns ratio: tight inductive coupling is provided between an input coil L_(p1) and secondary coil L_(s1). Tight inductive coupling is also provided between another primary coil L_(p2) and another secondary coil L_(s2). However, the L_(p1) /L_(s1) coils are not coupled at all to the L_(p2) /L_(s2) coils. This is preferably accomplished, using the integrated magnetic structure described above, by winding the L_(p2) and L_(s2) coils on one pole of a first flux circuit in a common physical magnetic structure, and winding the L_(p1) and L_(s1) coils, on another pole of the same structure, in a second flux circuit which is separate from the first flux circuit. The turns ratio of the L_(p1) /L_(s1) pair is preferably the same (N:1) as that of the L_(p2) /L_(s2) pair. Switch SW_(A) is operated to periodically pull one end of the input inductor L_(p2) toward ground. Capacitor C₁ ' provides coupling to the input winding L_(p1). Note that the two secondaries L_(s1) and L_(s2) are connected in parallel, in front of an output diode D_(F). This structure can reduce input ripple by steering ripple to the L_(p1) coil. However, maximal power efficiency and density are achieved by operating the two transformers at equal duty cycles, without ripple cancellation.

FIG. 3B is a timing diagram which shows the operation of the circuit of FIG. 3A. As may be seen from this timing diagram:

When the switch SW_(A) turns on:

the voltage V_(SW) across the switch goes to zero; the voltages V_(LP1) and V_(LP2) (on the two primary windings L_(P1) and L_(P2)) both jump up to V_(IN) ;

the voltages V_(LS1) and V_(LS2) (on the two secondary windings L_(s1) and L_(s2)) jump down to V_(IN) /N;

the current I_(LP1) on the input winding L_(P1) jumps up slightly, and then ramps up steadily;

the current I_(LP2) on the primary winding L_(P2) jumps up slightly (to zero), and then ramps up steadily; and

the currents I_(LS1) and I_(LS2) (on the two secondary windings L_(s1) and L_(s2))jump down to zero.

When the switch SWA turns off:

the voltage V_(SW) across the switch jumps up to V_(IN) +NV_(out) (with some overshoot due to the leakage inductance of L_(P1));

the voltages V_(LP1) and V_(LP2) (on the two primary windings L_(P1) and L_(P2)) both jump down to -NV_(out) (with some overshoot);

the voltages V_(LS1) and V_(LS2) (on the two secondary windings L_(s1) and L_(s2)) both jump up to V_(out) (with some overshoot);

the current I_(LP1) on the input winding L_(P1) jumps down to a substantially constant value I_(dc) ;

the current I_(LP2) on the primary winding L_(P2) jumps down to a substantially constant value -I_(dc) ;

the currents I_(LS1) and I_(LS2) (on the two secondary windings L_(s1) and L_(s2)) jump up, and then ramp down steadily.

Note that the two primaries L_(P1) and L_(P2) preferably transfer equal amounts of energy: the DC current -I_(dc) which flows in L_(P2) while the switch is off reduces the energy transferred into L_(s2), and the DC current I_(dc) which flows in L_(P1) while the switch is off increases the energy transferred into L_(s1), so that the peak current into L_(s2) is equal to the peak current into L_(s1).

Dual-Regenerative Flyback Converter

FIG. 4A shows a dual regenerative flyback converter circuit. This innovative topology differs from the isolated SEPIC topology in that two switches SW1 and SW2 are used. The use of two switches serves to tightly clamp the transient voltages which, as noted above, would otherwise appear (e.g. in the isolated SEPIC topology).

An input inductor L_(IN) is inductively coupled to a transformer L_(a) /L_(b). The turns ratio L_(IN) :L_(a) :L_(b) is N:N:1. Switches SW1 and SW2 close alternately; switch SW1 is connected from ground to the +side of capacitor C_(A), and switch SW2 is connected from ground to the- side of capacitors C_(A) and C_(B). The two switches SW1 and SW2 are preferably both power MOS devices of the same size. Note that switch SW1 is connected so that its parasitic diode prevents the + side of capacitor CA from going below the input ground, and switch SW2 (also preferably a VDMOS) is connected so that its parasitic diode prevents the--side of capacitors CA and C_(B) from going above the input ground. The energy in L_(a) can be discharged either to C_(B) or (through L_(b)) to the output capacitor C_(out). The volt-second balance in L_(IN) maintains the amp-second balance in C_(A).

FIG. 4B is a timing diagram which shows the operation of the circuit of FIG. 4A. Note that, in the presently preferred embodiment, switches SW1 and SW2 are operated in strict alternation.

When switch SW2 turns on and SW1 turns off:

the voltage V_(SW1) on switch SW1 jumps up to V_(IN) +NV_(out) (with virtually no ringing), and the voltage on switch SW2 drops to zero;

the voltage V_(LIN) on input inductor L_(IN), and the voltage V_(La) on primary inductor L_(a), both jump from V_(IN) to a negative value NV_(out) ;

the current I_(LIN) on input inductor L_(IN) ramps down at a rate of NV_(out) /L_(IN) ;

the current I_(La) on primary inductor L_(a) drops to zero;

the current I_(Lb) on secondary inductor L_(b) jumps up, and then ramps down.

When switch SW1 turns on and SW2 turns off:

the voltage V_(SW2) on switch SW2 jumps up to V_(IN) +NV_(out) (with virtually no ringing), and the voltage V_(SW1) on switch SW1 drops to zero;

the voltage V_(LIN) on input inductor L_(IN), and the voltage V_(La) on primary inductor L_(a), both jump up to V_(IN) ;

the current I_(LIN) on input inductor L_(IN) ramps up at rate of V_(IN) /L_(IN) ;

the current I_(La) on primary inductor L_(a) jumps up, and then ramps up; and

the current I_(Lb) on secondary inductor L_(b) drops to zero.

In the example shown the input inductor L_(IN) and transformer windings L_(a) /L_(b) are wound on separate flux circuits of a single planar magnetic structure like that of FIG. 1, so that input inductor LIN is not magnetically coupled to the transformer L_(a) /L_(b). However, in an alternative embodiment, inductive coupling can be added between the input inductor L_(IN) and the transformer windings L_(a) /L_(b), to steer ripple into the primary winding La and so minimize input ripple.

FIG. 4C shows another dual regenerative flyback converter circuit. This topology differs from that of FIG. 4A in that the input inductor L_(IN) ' is coupled to another inductor L₄ ' on the output side. This embodiment may be slightly slower than that of FIG. 4A to stabilize at startup, but otherwise retains many advantages. When this circuit topology is (advantageously) implemented with the magnetic structure of FIG. 1, the coils L₂ ' and L₃ ' are preferably wound on one pole of the core, and coils L_(IN) ' and L₄ ' are both wound on another pole of the core.

Innovative Portable Computer

FIG. 8 shows a portable computer including a power converter 800 as in FIGS. 3A, 4A, or 2A, which is used to charge the battery 802. The power converter is connected, through a full-wave bridge rectifier 120, to draw power from AC mains, and is connected to provide a DC voltage to the battery. The battery 802 (or the converter 800), connected through a voltage regulator 804, is able to power the complete portable computer system, which includes. in this example:

user input devices (e.g. keyboard 806 and mouse 808);

at least one microprocessor 810 which is operatively connected to receive inputs from said input device, through an interface manager chip 811 (which also provides an interface to the various ports);

memory (e.g. flash memory 812 and RAM 816), which is accesssible by the microprocessor;

a data output device (e.g. display 820 and display driver card 822) which is connected to output data generated by microprocessor; and

a magnetic disk drive 830 which is read-write accessible, through an interface unit 831, by the microprocessor 810.

Optionally, of course, many other components can be included, and this configuration is not definitive by any means.

According to a disclosed class of innovative embodiments, there is provided a power converter circuit comprising: an input winding operatively connected at a first end thereof to an input power terminal; an output winding inductively coupled to said input winding, and operatively connected to provide current at an output terminal; an intermediate capacitor and a primary winding, both connected in series with said input winding a switch connected to selectably vary the voltage of a node between said input and primary windings; and a secondary winding connected in parallel with said output winding, and inductively coupled to said primary winding but not to said input winding.

According to another disclosed class of innovative embodiments, there is provided a power converter circuit, comprising: an input winding operatively connected at a first end thereof to an input power terminal; a first intermediate capacitor, a second intermediate capacitor, and a primary winding, all jointly connected in series with said input winding; first and second switches connected to selectably vary the voltage of first and second nodes of one of said intermediate capacitors; and a secondary winding inductively coupled to said primary winding and to said input winding, and operatively connected to provide current at an output terminal.

According to another disclosed class of innovative embodiments, there is provided a circuit, comprising: an input winding operatively connected at a first end thereof to an input power terminal; an output winding inductively coupled to said input winding, and operatively connected to provide current at an output terminal; a first intermediate capacitor, a second intermediate capacitor, and a primary winding, all jointly connected in series with said input winding; first and second switches connected to selectably vary the voltage of first and second nodes of one of said intermediate capacitors; and a secondary winding connected in parallel with said output winding, and inductively coupled to said primary winding but not to said input winding.

Modifications and Variations

As will be recognized by those skilled in the art, the innovative concepts described in the present application can be modified and varied over a tremendous range of applications, and accordingly the scope of patented subject matter is not limited by any of the specific exemplary teachings given.

For example, as will be obvious to those of ordinary skill in the art, other circuit elements can be added to, or substituted into, the specific circuit topologies shown.

For another example, within the constraints well-known to those of ordinary skill, power MOS transistors can be replaced by IGBT and/or MCT devices, with appropriate allowance for reduced turn-off times. In some applications power bipolar devices can also be used.

For example, as will be obvious to those of ordinary skill in the art, other circuit elements can be added to, or substituted into, the specific circuit topologies shown.

For another example, within the constraints well-known to those of ordinary skill, nonlinear devices can be added in series with (or used to replace) resistors, to increase the impedance of load devices.

It should also be noted that the innovative circuit topologies of FIGS. 3A and 4A do not have to be used with the magnetic structure of FIG. 1, although this combination is believed to be particularly advantageous.

Additional discussion of alternatives and implementation details known to those skilled in the art can be found in the publications cited in the Background section of the present application, and in the following publications, all of which are hereby incorporated by reference: Billings, SWITCHMODE POWER SUPPLY HANDBOOK (1989); Chetty, SWITCH-MODE POWER SUPPLY DESIGN (1986); Chryssis, HIGH FREQUENCY SWITCHING POWER SUPPLIES (2.ed. 1989); Flanagan, HANDBOOK OF TRANSFORMER DESIGN & APPLICATIONS (2.ed. 1993); Gottlieb, POWER SUPPLIES, SWITCHING REGULATORS, INVERTERS, AND CONVERTERS (2.ed. 1994); Holt, SEMICONDUCTOR POWER ELECTRONICS (1986); Lenk, SIMPLIFIED DESIGN OF SWITCHING POWER SUPPLIES (1995); Mazda, POWER ELECTRONICS HANDBOOK (1990); Mohan et al., POWER ELECTRONICS (2.ed. 1995); Nasar, ELECTRIC MACHINES AND TRANSFORMERS (1984); Nave, POWER LINE FILTER DESIGN FOR SWITCHED-MODE POWER SUPPLIES (1991); REACTIVE POWER: BASICS, PROBLEMS AND SOLUTIONS (ed. Sheble 1987); Severns and Bloom, MODERN DC-TO-DC SWITCHMODE POWER CONVERTER CIRCUITS (1984); Shepard, POWER SUPPLIES (1984); Sum, SWITCH S MODE POWER CONVERSION (1988); Tihanyi, ELECTROMAGNETIC COMPATIBILITY IN POWER ELECTRONICS (1995); Williams, POWER ELECTRONICS (1987); Wood, SWITCHING POWER CONVERTERS (1981); the proceedings of the annual INTERNATIONAL HIGH-FREQUENCY POWER CONVERSION conferences from 1986 to date; and the proceedings of the POWERCON and POWER ELECTRONICS SPECIALISTS conferences from 1980 to date. All of these books, and the references cited in them, are hereby incorporated by reference. 

What is claimed is:
 1. A power converter circuit comprising:an input winding operatively connected at a first end thereof to an input power terminal; an output winding inductively coupled to said input winding, and operatively connected to provide current at an output terminal; an intermediate capacitor and a primary winding, both connected in series with said input winding; switch connected to selectably vary the voltage of a node between said input and primary windings; and secondary winding connected in parallel with said output winding, and inductively coupled to said primary winding but not to said input winding.
 2. The circuit of claim 1, wherein all said windings are wound on a low-profile integrated magnetic structure which includes multiple separate flux paths, and some of said windings are decoupled from others of said windings.
 3. The circuit of claim 1, wherein all said windings are wound on a common magnetic structure.
 4. The circuit of claim 1, wherein said power output terminals are configured to provide power to a computer system.
 5. The circuit of claim 1, wherein said power output terminals are configured to provide power to an electrical storage and recharging system.
 6. The circuit of claim 1, wherein said switch is a VDMOS power transistor.
 7. A power converter circuit, comprising:an input winding operatively connected at a first end thereof to an input power terminal; a first intermediate capacitor, a second intermediate capacitor, and a primary winding, all jointly connected in series with said input winding; first and second switches connected to selectably vary the voltage of first and second nodes of one of said intermediate capacitors; and a secondary winding inductively coupled to said primary winding and to said input winding, and operatively connected to provide current at an output terminal.
 8. The circuit of claim 7, wherein all said windings are wound on a low-profile integrated magnetic structure which includes multiple separate flux paths, and some of said windings are decoupled from others of said windings.
 9. The circuit of claim 7, wherein said power output terminals are connected to provide power to a portable computer system.
 10. The circuit of claim 7, wherein said power output terminals are operatively connected to charge a battery.
 11. The circuit of claim 7, wherein at least one said capacitor is electrolytic.
 12. The circuit of claim 7, wherein all said windings are wound on a common magnetic structure.
 13. The circuit of claim 7, wherein each said switch is a VDMOS power transistor.
 14. A circuit, comprising:an input winding operatively connected at a first end thereof to an input power terminal; an output winding inductively coupled to said input winding, and operatively connected to provide current at an output terminal; a first intermediate capacitor, a second intermediate capacitor, and a primary winding, all jointly connected in series with said input winding; first and second switches connected to selectably vary the voltage of first and second nodes of one of said intermediate capacitors; and a secondary winding connected in parallel with said output winding, and inductively coupled to said primary winding but not to said input winding.
 15. The circuit of claim 14, wherein all said windings are wound on a low-profile integrated magnetic structure which includes multiple separate flux paths, and some of said windings are decoupled from others of said windings.
 16. The circuit of claim 14, wherein said power output terminals are connected to provide power to a portable computer system.
 17. The circuit of claim 14, wherein said power output terminals are operatively connected to charge a battery.
 18. The circuit of claim 14, wherein all said windings are wound on a common magnetic structure.
 19. The circuit of claim 14, wherein at least one said capacitor is electrolytic.
 20. The circuit of claim 14, wherein said switch is a VDMOS power transistor. 